The need for high-frequency power conversion has prompted research in quasi-resonant converters employing zero-current and zero-voltage-switching techniques.
Zero-current-switched quasi-resonant converters (ZCS-QRCs) reduce turn-off losses by shaping the switching transistor current to zero prior to turn-off. This allows ZCS-QRCs to operate at frequencies up to about 2 MHz. Further increase of the switching frequency of ZCS-QRCs is difficult to accomplish because of capacitive turn-on loss. Also, the Miller effect comes into play in that it relates to turn-on of the transistor at non-zero-voltage and the resultant parasitic oscillations caused by the output capacitance of the transistor.
See U.S. Pat. No. 4,720,667 (Lee et al) for several examples of zero-current-switched quasi-resonant converters.
Zero-voltage-switched quasi-resonant converters reduce the problem of turn-off losses by shaping the switching transistor voltage to zero prior to turn-on. As a result, ZVS-QRCs can operate at higher frequencies, up to 10 MHz. However, the ZVS-QRCs have two major limitations. One problem is excessive voltage stress to the switching transistor proportional to the load range. This makes it difficult to implement ZVS-QRCs with wide load variations. Another problem is caused by the junction capacitance of the rectifying diode used in the quasi-resonant converter. When the diode turns off, this junction capacitance oscillates with the resonant inductance. If damped, these oscillations cause significant power dissipation at high frequencies; undamped, they adversely affect the voltage gain of the quasi-resonant converter and, thus, the stability of the closed-loop system.
See U.S. Pat. No. 4,720,668 (Lee et al) for several examples of zero-voltage-switched quasi-resonant converters.
FIGS. 1a and 1b show the equivalent circuits of prior art zero-current and zero-voltage quasi-resonant switches. Each of these topologies represents a high-frequency sub-circuit extracted from a quasi-resonant converter by replacing voltage sources and filter capacitors with short circuits and filter inductors with open circuits. In the equivalent circuit of the zero-current quasi-resonant switch, shown in FIG. 1a, the active switch S operates in series with the resonant inductor L while the diode D operates in parallel with the resonant capacitor C.sub.D. In the zero-voltage quasi-resonant switch shown in FIG. 1b, the situation is opposite. The active switch S is in parallel with the capacitor C.sub.S and the diode D is in series with the inductor L. It can be easily seen that the two topologies are dual.
FIG. 2 shows the circuit diagram of a buck ZCS-QRC. This topology is derived from a pulse width modulation (PWM) buck converter by inserting a resonant inductor L in series with the switch S and a resonant capacitor C.sub.D in parallel with the diode D. Anti-parallel diode D.sub.S represents the body diode of a metal oxide semiconductor field effect transistor (MOSFET) which is typically used as the high-frequency switch S. The filter in the output stage is formed by inductor L.sub.F and capacitor C.sub.F. Resistor R.sub.L represents the load.
When switch S is conducting, inductor L and capacitor C.sub.D resonate. Current through switch S is sinusoidal and reduces to zero before switch S is turned off. This, in theory, eliminates losses related to inductive turn-off. In practice, however, reverse recovery of the body diode of the MOSFET causes harmful oscillation between the resonant inductor and output capacitance of the MOSFET. To avoid this oscillation, a diode is added in series with switch S to prevent the current from flowing into diode D.sub.S. The resulting half-wave mode of operation not only increases conduction losses, but also makes the converter load-sensitive. The minimum switching frequency at light load is reduced substantially and leads to larger filter components and slower transient response.
Although ZCS-QRCs take advantage of zero-current turn-off, turn-on occurs when full input voltage is applied to the switch. This causes dissipation of the energy stored in the output capacitance of the switch and change in voltage per unit time (dv/dt) noise which is coupled through the drain-to-gate capacitance of the power MOSFET to the gate-drive circuit (switching Miller effect).
In ZCS-QRCs, the switching conditions for the active device are not of the most favorable variety. However, switching conditions for the rectifying diode on the other hand, are very favorable. The reason for this is that power diodes are easy to turn on, but the reverse recovery characteristics of such devices often result in excessive turn-off loss and noise. The most favorable condition to turn off a diode occurs when current reduces gradually to zero and no immediate reverse voltage is applied to the diode afterwards. This is the case for ZCS-QRCs. Again, with reference to FIG. 2, when switch S is turned on, the current through diode D decreases linearly until it reaches zero. Then the diode turns off and the voltage across it builds up gradually in a resonant fashion. The only disadvantage is that the reverse voltage applied across the diode is approximately twice the input voltage. The maximum switching frequency of ZCS-QRCs is limited due to the turn-on switching loss in the active switch.
The ZVS-QRC topology, shown in FIG. 3, is derived from its pulse-width-modulation (PWM) counterpart by adding a resonant capacitor C.sub.S in parallel with the switch S and a resonant inductor L in series with the diode D. The inductor can be placed anywhere in the resonant loop provided the resonant switch, extracted from the circuit, is always reduced to the topology of FIG. 1b. in the ZVS-QRC of FIG. 3, the active switch operates under favorable switching conditions. At turn-off, the current is diverted from the switch into the resonant capacitor C.sub.S which is, subsequently, being charged linearly to the input voltage by the load current flowing through L and L.sub.F. The gradual increase of V.sub.S minimizes overlapping of the switch current and voltage at turn-off, thus, reducing the switching losses. The turn-on condition is even better, since the voltage across the switch resonates and reduces to zero prior to turn-on. This turn-on condition totally eliminates the capacitive turn-on losses and the switching Miller effect associated with ZCS-QRCs.
Improved switching conditions for the active switch S allow ZVS-QRCs to operate at 10 MHz. However, the operation of ZVS-QRCs is adversely affected by the undesired switching conditions created for the rectifying diode. In particular, immediately after the diode current reduces to zero, voltage applied to the diode changes abruptly from zero to V.sub.IN. Such an abrupt voltage change induces parasitic oscillations between the resonant inductor and diode capacitance. During conduction of switch S, the current through the switch and voltage across the diode are oscillatory. In practice, these oscillations typically do not decay before switch S is turned off, as shown in FIGS. 4a and 4b, which respectively show the theoretical and experimental voltage waveform of the rectifying diode in the ZVC-QRC of the type shown in FIG. 3. The undesired oscillation adversely affects the conversation ratio characteristics. FIGS. 5a and 5b are waveforms illustrating the effect of parasitic junction capacitance C.sub.j of the rectifying diode on the DC conversion ratio characteristics of the buck ZVS-QRC of FIG. 3. FIG. 5a shows the ideal characteristics at C.sub.j =0 and FIG. 5b shows the characteristics at C.sub.j =0.5 C.sub.S.
In each graph, M represents the conversion ratio V.sub.O /V.sub.IN and f.sub.N represents the normalized frequency f/f.sub.S, where f.sub.S =1/2.pi..sqroot.LC.sub.S, with normalized output current, I.sub.N being equal to I.sub.O .sqroot.L/C.sub.S /V.sub.IN as a free running parameter. When the junction capacitance C.sub.j of the rectifying diode D.sub.S is assumed to be zero, the characteristics are shown as straight lines in FIG. 5a. FIG. 5b shows characteristics for C.sub.j =0.5 C.sub.S. It can be seen that even if junction capacitance C.sub.j is only half of the resonant capacitance C.sub.S, its effects are quite pronounced, for example, in high-frequency converters, the junction capacitance C.sub.j can easily be larger than C.sub.S, especially if high-current diodes with large die areas are used. The discontinuity of the characteristics implies that the zero-voltage-switching property is lost for some operating conditions. Furthermore, in regions where the slopes of the curves are positive, the converter exhibits local closed-loop instabilities. Even in those regions where the slopes are negative, the slope can be very steep which makes the conversion ratio very sensitive to the switching frequency and, thus, difficult to control.
Another important concern of ZVS-QRCs is extensive voltage stress at the switching transitor. Typically, this stress is proportional to the load range. For example, in a buck ZVS-QRC, this stress is V.sub.Smax =V.sub.IN (1+R.sub.Lmax /R.sub.Lmin). Thus, for a load range of 10:1, voltage stress is 11 times the input voltage.
In the above discussion, it has been shown the zero-current-switched and zero-voltage-switched quasi-resonant techniques optimize switching conditions for either the active switch or the diode, but not for both simultaneously. Furthermore, each application is limited primarily by the undesired parasitic oscillations in the circuit. The ZCS-QRCs are adversely affected by the body diode and output capacitance of the power MOSFET, while the ZVS-QRCs deteriorate due to the junction capacitance of the rectifying diode.
There is thus a need for a resonant switching network that operates under switching conditions that are favorable to both the active switch and the diode. The present invention is directed toward filling that need.